High performance class ab operational amplifier

ABSTRACT

A class AB operational amplifier includes an input stage, an output stage and a level shifter stage to control the quiescent current of the output stage and to transfer the signal from the input stage to the output stage, and a control circuit of the level shifter stage. The control circuit includes a transistor differential pair having a differential input terminals and the differential voltage at the differential terminals of the differential pair controls the level shifter stage.

RELATED APPLICATIONS

This application is a translation of and claims the priority benefit of Italian patent application number MI2011A001832, filed on Oct. 7, 2011, entitled High Performance Class AB Operational Amplifier, which is hereby incorporated by reference to the maximum extent allowable by law.

FIELD OF THE INVENTION

The present invention relates to a high performance class AB operational amplifier. The design of high-accuracy analogue circuits is becoming a difficult task with the scaling down of supply voltages and transistor channel lengths of the current mixed-signal integrated circuits. Most of these circuits require the use of high performance active cell: the operational amplifier (OpAmp). Furthermore, some low-IF transceivers make use of complex filters which require operational amplifier with a very high gain-bandwidth product in order to have enough loop gain at the higher band limit. Ultra low power circuit imposes a current reduction in order to increase the battery life-time in mobile handsets. Unfortunately, the bandwidth depends on the technology and the current used in the OpAmp output stage and hence the only way to reduce the current consumption is to reduce the complexity/consumption of the circuits around the OpAmp such as bias, common mode feed-back, start-up circuit. Designers are continuously working toward tradeoff solutions between gain, input/output swings, speed, power consumption, noise, etc.

The class-AB output stage topology allows driving a large load capacitor with a small output stage bias current. Furthermore, it gives a boost in the Gain Bandwidth product (GBW). Despite to its property, the most important problem designing class AB OpAmps is the accurate control of the output current which depends on process and supply voltage variations. To solve this problem it is necessary to introduce control circuits.

An ideal circuit to implement a class AB output stage is shown in FIG. 1. The circuit comprises an input stage comprising a differential pair of PMOS transistors M_(1a), M_(1b) having a differential input signal at the input terminals IN_(P), IN_(M). The drain terminals of the transistor of the differential pair are connected to the output terminals OUT_(P), OUT_(M) by means of the compensation network constituted of a series of a capacitor C_(c) and a resistance R_(c). The output terminals OUT_(P), OUT_(M) belonging to two output stages each one comprising a couple of transistor PMOS M_(7a), M_(7b) and NMOS M_(6a), M_(6b) having the drain terminal in common and connected to the output terminals OUT_(P), OUT_(M) and the two level shifter Δv/2 connected between the output terminals of the differential pair and the gate terminals of the transistor PMOS M_(7a), M_(7b) and NMOS M_(6a), M_(6b) of the couples.

The biasing of the differential pair is performed by means of the transistors NMOS M_(2a), M_(2b) having the gate terminal connected to the bias voltage VB_(N) and the source terminals of the differential pair transistors connected to the drain terminal of a PMOS transistor M₃ having the gate terminal connected to a bias voltage VB_(p) and the source terminal connected to the supply voltage Vdd as the source terminals of the transistors M_(7a), M_(7b); in this way a bias current Iss/2 flows in each branch of the differential pair.

The circuit in FIG. 1 allows to control the bias current and transfer the signal to both the NMOS and PMOS transistor of the output stage using two level shifter Δv/2.

A practical implementation of the circuit in FIG. 1 is shown in FIG. 2. The two level shifter Δv/2 of the output stages are implemented by means of a PMOS and a NMOS transistor arranged in series M_(4a), M_(5a) and M_(4b), M_(5b) and connected between the supply voltage Vdd and ground (GND). The gate terminals of the transistors M_(5a), M_(5b) are connected to the output terminals of the differential pair while the gate terminals of the transistors M_(4a), M_(4b) are connected to the bias voltage VB_(p).

The circuit topology in FIG. 2 assures the best performance in terms of GBW, power consumption and output voltage swing. Unfortunately, this topology has many drawbacks to be solved; the output-stage current is not controlled and depends on the process, mismatch of the MOS transistors and on the supply voltage. Furthermore, this topology is not at low voltage. The minimum supply voltage which can be used depends on the MOS transistors threshold V_(TH) and is V_(THN)+2V_(THP) where V_(THN) is the threshold of the NMOS transistor while V_(THP) is the threshold of the PMOS transistor.

Another circuit topology used is shown in FIG. 3. This circuit solution comprises the use of a circuit 1 to control the output current Tout of the class AB output stage in FIG. 2; FIG. 3 shows only a part of the circuit in FIG. 2 which is relative to the input IN_(M) and the output OUT_(M). The gate terminal of the transistor M₄ is not connected with the bias voltage VB_(P) but with the circuit 1, more precisely the gate terminal of the transistor M₄ is connected with the gate terminal and the drain terminal of a PMOS transistor M_(4r) and with the gate terminal of a PMOS transistor M₃, so that the transistor M_(4r) and M_(3r) form a current mirror. Another current mirror comprises the transistor M_(1r) and M_(6r) adapted to mirror the bias current I_(B) in the PMOS transistor M_(7r) having the gate terminal connected with the drain terminal of the transistor M_(3r) and with the source terminal of a PMOS transistor M_(5r) the drain terminal of which is connected to ground GND and the gate terminal is connected with the gate terminal of the transistors M_(1r) and M_(6r). Loop consists of transistors M_(3r), M_(7r), M_(2r) and M_(4r). The gate-source voltage of the transistor M_(5r) will be adjusted corresponding to the currents flowing through the transistors M_(7r) and M_(6r) The current flowing through the transistor M_(5r) is copied in the transistor M₄ by the transistor M_(3r) and the gate-source voltage of the transistor M₅ is adjusted to a right level forcing the output stage current to the controlled level.

The circuit in FIG. 3 is known from: H. A. Aslanzadeh, S. Mehrmanesh, M. B. Vahidfar, A. Q. Safarian, “A low power 25 Ms/s 12-bit pipelined analog to digital converter for wireless applications,” IEEE 2003.

This solution is not suitable for very low current consumption application; in the case of low bias current the output current depends on the process variation of the transistor and the performances of the class AB output stage of the operational amplifier are reduced.

In view of the state of the art, the object of the present invention is to provide a high performance class AB operational amplifier which overcomes the above-mentioned drawbacks.

SUMMARY OF THE INVENTION

According to the present invention, this object is achieved by means of a class AB operational amplifier comprising an input stage, an output stage and a level shifter stage adapted to control the quiescent current of the output stage and to transfer the signal from the input stage to the output stage, a control circuit of the level shifter stage, characterized in that the control circuit comprises at least one transistor differential pair having differential input terminals, said control circuit being configured so that the differential voltage at the differential terminals of the transistor differential pair controls the level shifter stage.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and advantages of the present invention will become apparent from the following detailed description of an embodiment thereof, illustrated only by way of non-limitative example in the annexed drawings, in which:

FIG. 1 shows an ideal circuit scheme of a class AB operational amplifier output stage;

FIG. 2 shows a circuit implementation of the circuit scheme in FIG. 1;

FIG. 3 shows a circuit implementation of the circuit scheme in FIG. 1;

FIG. 4 shows a class AB operational amplifier according to the present invention;

FIGS. 5 and 6 are time diagrams of the gain and phase of the circuit in FIG. 4 deriving from simulations; and

FIG. 7 shows montecarlo simulations of the output stage current of the class AB operational amplifier in FIG. 4.

DETAILED DESCRIPTION

FIG. 4 shows a class AB operational amplifier according to the present invention. The operational amplifier comprising an input stage 10, an output stage 11, a level shifter stage 12 adapted to control the bias current of the output stage and to transfer the signal from the input stage to the output stage, a control circuit 13 of the level shifter stage. Specifically, FIG. 4 shows one part of the class AB operational amplifier according to the present invention which comprises two symmetrical parts for each branch of the input stage that is two parts identical to each other.

The input stage 10 comprising a differential pair of transistors M_(1a), M_(1b), preferably PMOS transistor, having a differential input signal at the input terminals IN_(P), IN_(M) and a common node. The input stage 10, similar to the input stage 1 in FIG. 1, has a bias current Iss/2 flowing through the transistor M_(1a), M_(1b).

The output stage 11 comprises a couple of a first M₆ and a second M₇ output transistors, in series to each other, and arranged between a first voltage reference Vdd and a second voltage reference GND and being interconnected at an output terminal OUT of the operational amplifier; the first output transistor has a control terminal connected to an output terminal of the input stage 10.

The level shifter stage 12 comprises a couple of a first M₅ and a second M₉ intermediate transistors, connected in series to each other by means of an impedance, that is the parallel of a resistance R and a capacitor C, and coupled between a first voltage reference Vdd and a second voltage reference GND; specifically, the transistors of the level shifter stage 12 are MOS transistors and the source terminal A of the first transistor M₅ is coupled with the drain terminal B of the second transistor M₉ by means of the parallel of the resistance R and the capacitor C. The first intermediate transistor M₅ has the control terminal connected with one output terminal of the input stage 10 and the second intermediate transistor M₉ has the output terminal connected with the control terminal of the second output transistor M₇; the second intermediate transistor M₉ has the control terminal connected to an output terminal of a control circuit 13. The second intermediate transistor acts as a current generator.

The control circuit 13 comprises at least one transistor differential pair M_(8a), M_(8b) having a differential input terminals B′ and A′. The differential voltage V_(A′)−V_(B′), applied at the input terminals A′ and B′ controls the level shifter stage 12. In fact the level shifter stage 12 controls the output quiescent current lout of the output stage 11 of the operational amplifier by means of the voltage difference between the voltages at the terminals A and B; the control circuit 13 is configured so that the differential voltage V_(A′)−V_(B′). is equal to the differential voltage V_(A)−V_(B) applied at the terminals A and B. Particularly, the level shifter stage 12 fixes the voltage V_(A) while the voltage V_(B) is varied by the control circuit 13.

The control circuit 13 comprises replica circuits based on transistors M_(7r), M_(6r) and M_(5r). The term “replica” is used herein to refer to a copy or duplicate of an original, which may be of different scale to the original. Preferably, the transistors M_(7r), M_(6r) and M_(5r) are a scaled version of transistors M₇, M₆ and M₅ respectively. The scaling value depends on the current ratio desired between transistors M₅ and M_(5r)=M₅/k, M₆ and M_(6r)=M₆/n, M₇ and M_(7r)=M₇/n. This ratio of current in the stages 10, 11 and 12 and in the replica-circuit branches is configured to reduce the current consumption of the control circuit 13.

The control circuit 13 comprises circuit branches 14, 15 connected with the input terminals B′, A′ of the differential pair M_(8a), M_(8b); the circuit branches 14, 15 are replica circuits of the respective circuit branches which are connected with the respective terminals B and A and which are formed by the transistor M₇ and M₅, M₆ among the output stage 11 and the level shifter stage 12.

The currents of the transistors M_(7r), M_(6r), M_(6r) and M_(5r) are fixed through current sources to a scaled value of the desired current flowing through the transistors M₇, M₆ and M₅. This allows to evaluate the desired voltage on the node A and B with process and temperature variations. The evaluated value is available at the node A′ and B′.

The control circuit 13 comprises first and second circuit branches 14, 15 respectively formed by the transistor M_(7r) wherein the bias current Ioutr flows, and by the transistor M_(6r) wherein the bias current Ioutr flows and the transistor M_(5r) having the gate terminal connected with the gate terminal and with the drain terminal of the transistor M_(6r). The drain and source terminals of the transistors M_(7r), M_(5r) are connected to the terminals of the differential pair M_(8a), M_(8b) so as the differential voltage V_(A′)−V_(B′) is proportional to the bias current Ioutr.

A transistor M_(9r) forms a mirror with the transistor M₉ of the level shifter stage. The transistor M_(9r) has the gate and the drain terminals connected with the drain terminal of the transistor M_(8a); another transistor M_(9r) has the gate and the drain terminals connected only with the drain terminal of the transistor M_(8b).

The differential pair M_(8a), M_(8b) compares the voltages V_(A′). and V_(B′) and change the value of current in the transistor M_(9r) from I_(Br) to the value I_(Br)−ΔI with ΔI=(V_(A′)−V_(B′))/R_(r). This current is mirrored in the transistor M₉ with a mirror factor k. The current flowing through M₉ is k*(I_(Br)+ΔI). If the currents I_(B) and I_(Br) are related by I_(B)=k•I_(Br), an extra current k•ΔI flows through the resistor R and produces a variation k•ΔI•R between the voltage of the node A and B, that is V_(A)−V_(B)=k•ΔI•R=k•R•(V_(A′)−V_(B′))/R_(r). If the relation chosen between R and R_(r) is R_(r)=k•R then it has V_(A)−V_(B)=V_(A′)−V_(B′); the control circuit 13 guarantees at the node A and B the voltage which is necessary in order to have the desired current Iout=n•Ioutr in the output stage M₇−M₆ with process and temperature variations, that is each variation of the current Ioutr determines a variation of the current lout of the output stage 11 of the operational amplifier in class AB.

Since the transistors M_(8a), M_(8b) are configured as follower transistors, the voltage difference at the terminals of the resistance R_(r) is the voltage difference V_(A)′−V_(B′). The voltage V_(A′) is given by the sum of the gate-source voltages of the transistors M_(6r) and M_(5r), that is

$V_{A^{\prime}} = {{V_{T_{p}}} + \sqrt{\frac{I_{Br}}{{K_{p}\left( \frac{W}{L} \right)}_{5r}}} + V_{Tn} + \sqrt{\frac{I_{outr}}{{K_{n}\left( \frac{W}{L} \right)}_{6\; r}}}}$

wherein V_(Tp) is the voltage threshold to the PMOS transistor, V_(Tn) is the voltage threshold to the NMOS transistor, W/L is the form factor and K_(p) and K_(n) are constants. The voltage V_(B′) is given by the difference between the supply voltage Vdd and the gate-source voltage of the transistor M_(7r), that is

$V_{B^{\prime}} = {{Vdd} - {V_{Tp}} - {\sqrt{\frac{I_{outr}}{{K_{n}\left( \frac{W}{L} \right)}_{7\; r}}}.}}$

The voltage V_(A) is given by the sum of the gate-source voltages of the transistors M₆ and M₅, that is

$V_{A^{\prime}} = {V_{T_{N}} + \sqrt{\frac{I_{out}}{{K_{n}\left( \frac{W}{L} \right)}_{6}}} + {V_{Tp}} + \sqrt{\frac{I_{B}}{{K_{p}\left( \frac{W}{L} \right)}_{5}}}}$

and the voltage V_(B) is given by the difference between the supply voltage Vdd and the gate-source voltage of the transistor M₇, that is

$V_{B^{\prime}} = {{Vdd} - {V_{Tp}} - {\sqrt{\frac{I_{out}}{{K_{p}\left( \frac{W}{L} \right)}_{7}}}.}}$

Since V_(A)−V_(B)=V_(A′)−V_(B′) with R_(r)=k•R and I_(B)=k•I_(Br) by selecting

$\left( \frac{W}{L} \right)_{5} = {{{k\left( \frac{W}{L} \right)}_{5r^{\prime}}\left( \frac{W}{L} \right)_{6}} = {{{n\left( \frac{W}{L} \right)}_{6r}\mspace{14mu} {and}\mspace{20mu} \left( \frac{W}{L} \right)_{7}} = {n\left( \frac{W}{L} \right)}_{7r}}}$

it occurs lout=n×loutr.

This configuration has the advantage to eliminate the dependency of the output stage quiescent current on transistor process variation and supply voltage while controlling this current in an open loop manner which does not affect the frequency response and stability of the amplifier.

FIGS. 5 and 6 show the differential-mode open-loop gain G in decibel and the phase in degrees of the operation amplifier in FIG. 4 when it is connected in unity gain feed-back with a current consumption of 150 microampere, a power supply of 1.2 V and a load capacitor of 100 fF single-ended. These curves show a perfect two pole system. The operational amplifier exhibits a DC gain G of 50 dB and a unity-gain frequency of 500 MHz and 60 degree of phase margin.

FIG. 7 shows the montecarlo simulations with process and temperature variation; in said simulations the number of cases n is in ordinate and the current lout (in microampere) is in abscissa, with the total number of cases equal to 100, at 100° C., 27° C. and −40° C. The montecarlo simulations show a very accurate control of the output-stage current.

As above mentioned FIG. 4 shows one part of the class AB operational amplifier according to the present invention which comprises two symmetrical parts for each branch of the input stage, that is two parts identical to each other and wherein each one of two parts includes the circuits 11, 12 and 13 shown in FIG. 4.

While the invention has been described by way of example and in terms of the preferred embodiments, it is to be understood that the invention is not limited to the disclosed embodiments. To the contrary, it is intended to cover various modifications and similar arrangements (as would be apparent to those skilled in the art). Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications and similar arrangements. 

1. A class AB operational amplifier comprising an input stage, an output stage and a level shifter stage adapted to control the quiescent current of the output stage and to transfer the signal from the input stage to the output stage, a control circuit of the level shifter stage, characterized in that the control circuit comprises at least one transistor differential pair having differential input terminals, said control circuit being configured so that the differential voltage at the differential terminals of the transistor differential pair controls the level shifter stage.
 2. The operational amplifier according to claim 1, characterized in that said control circuit comprises circuit means coupled with the input terminals of the transistor differential pair and wherein a first bias current flows, the control circuit being configured to control the quiescent current to be proportional to the amount of said first bias current.
 3. The operational amplifier according to claim 2, characterized in that the level shifter stage comprises at least a first and a second intermediate transistors coupled in series by means of an impedance and the series being arranged between a first and a second reference voltage, the voltage across the impedance being said differential voltage at the differential input terminals of the transistor differential pair.
 4. The operational amplifier according to claim 3, characterized in that said circuit means of the control circuit comprises a first and a second circuit branches connected with the respective input terminals of the differential pair, said circuit branches being a replica circuit of the two circuit branches of the output stage and the level shifter stage which are connected at the respective terminals of the impedance.
 5. The operational amplifier according to claim 3, characterized in that said transistor differential pair comprises MOS transistor and a resistance arranged between the source terminals of the differential pair, said impedance comprising a further resistance and said resistance being proportional to the further resistance, said transistor differential pair being configured so that the voltage across the resistance is the differential voltage at the differential terminals of the differential pair.
 6. The operational amplifier according to claim 5, characterized in that said control circuit comprises a first transistor forming with an intermediate transistor of the level shifter stage a current minor to mirror the current flowing through the transistor differential pair into the level shifter stage, said first transistor being a replica of said intermediate transistor.
 7. The class AB operational amplifier of claim 1 further comprising an additional input stage, output stage, level shifter stage, and control circuit.
 8. A class AB operational amplifier comprising: an input stage; an output stage; a level shifter stage to control the bias current of the output stage and to transfer the signal from the input stage to the output stage; and a control circuit of the level shifter stage.
 9. The class AB operational amplifier of claim 8 wherein the input stage comprises a differential pair of transistors.
 10. The class AB operational amplifier of claim 9 wherein the transistors comprise PMOS transistors.
 11. The class AB operational amplifier of claim 8 wherein the output stage comprises a first transistor and a second transistor, in series connection.
 12. The class AB operational amplifier of claim 11 wherein the first transistor comprises a PMOS transistor and the second transistor comprises an NMOS transistor.
 13. The class AB operational amplifier of claim 11 wherein a series-connected resistor and capacitor are coupled between a gate and a drain of the second transistor.
 14. The class AB operational amplifier of claim 8 wherein the level shifter stage comprises a first transistor and a second transistor, coupled together by an impedance.
 15. The class AB operational amplifier of claim 14 wherein the first transistor comprises a PMOS transistor and the second transistor comprises an NMOS transistor.
 16. The class AB operational amplifier of claim 14 wherein the impedance comprises a resistor and a capacitor in parallel.
 17. The class AB operational amplifier of claim 14 wherein an intermediate node between the first transistor and the impedance is coupled to ground.
 18. The class AB operational amplifier of claim 8 wherein the control circuit 13 comprises a transistor differential pair.
 19. The class AB operational amplifier of claim 18 further comprising a first circuit branch and a second circuit branch coupled to the transistor differential pair.
 20. A class AB operational amplifier comprising: two symmetrical circuit portions, each circuit portion comprising: an input stage; an output stage; a level shifter stage to control the bias current of the output stage and to transfer the signal from the input stage to the output stage; and a control circuit of the level shifter stage. 